The markers at approximately
(wanted 2nd IF) and 33kHz (quantisation aliasing frequency) show how
little rolloff the original circuit has. Maybe the deep notch around
90kHz was intended to be at the alias frequency?
In the schematic below, I have
shown the PI-pad values that I used; the graph above uses the
standard values which give 3dB more attenuation.
This is a part-way solution
and is not final.
Note that I have changed the coupling capacitors from the original (too
small) values here, since this is required anyway. Although the scales
for the responses have altered, it can be seen that
distributing the 50R HF load between the 2 arms of the filter (R10, R11
above) results in a deeper trough due to the higher Q of the resonant
series arm. These two configurations are the best I could manage with
this topology. Note that the change to a high value of R1 (47k) in the
above schematics is disabling the original HF arm of the diplexer. The
HF termination is done with the new 47R (R10) or with the series
combination (R10, R11).
The attenuation at 33kHz relative to 15kHz is approximately:
- Cauer-2 configuration, 7dB
- Cauer-1 configuration, 11dB (in spite of the shallower trough).
- Simple filter for comparison, 9dB.
So I conclude that the Cauer, once sanitized not to shunt the output
capacitively at higher frequencies, does not provide any great benefit
over the simpler, conventional 'L' filter.
In any event, we must make the identical circuit serve in the Transmit
After studying the Transmit Path, it seemed a good idea to use a 'T'
low-pass filter rather than an 'L' configuration, mainly in order to
capacitive load on the transmit-side buffer. The 'modified Cauer'
(Cauer-2) above almost gets there, by presenting a 22R resistive load
at high frequencies, but there are other possibilities.
When studying across sheets of the schematics and drawing a scrap
circuit of things, it became obvious that
the output of the MixAdaptor section is biassed at 2.5V, so the
diplexer receives this d.c. offset together with the signal. Then,
following the filter, there is a capacitor to 'isolate' this d.c.
component, followed immediately by reinstating it in order to bias the
switch IC604. The implication is that there is no voltage
across C632, so I measured it: 12mV! There
alone a 10uF-value C632; it can be replaced with a short-circuit (and
bias divider R625, R626 removed or not fitted if desired).
With this in mind, I considered replacing C632 with an inductor rather
than a short-circuit, thereby building a 'T' low-pass filter consisting
RFC612, C651 and the 'C632' inductor. This would allow not only the
emitters of TR604, TR606 to see a load that increases at high
but also gives the filter another element. However, on looking
for inductors of 'reasonable' value, hundreds of uH, in a surface-mount
package small enough to fit in place of C632, it became apparent that
they don't come this physically small. One thing we are avoiding is
Undeterred and raring to see if it was worth trying, I attempted to
design a 'T' filter that would have something like 50R looking both
and out of it, a symmetrical filter, but with the bias towards the
receive direction being good, since on Rx we are trying to pick signals
out of received -um- stuff. On Tx the signal from the DSP is as good as
it can be. After a good bit of head-scratching
and nearly wearing out Elsie <g>, I tried some 'handy' values and
hit gold! By using a larger value for C651, 200nF (it was 100nF
for the 'L' filter) and adding 330uH in place of C631, I ended up with
a filter in the receive direction with a sharper cutoff than before,
one in the
transmit direction with a reasonably respectable behaviour! Too
good to ignore, I discovered that the axial 330uH chokes from Farnell
would fit the
surface mount footprint if the legs are bent appropriately! The body
stands vertically, but who cares? It's a bit tight for space, because
of adjacent components, but it fits (note that Glenn later used a
different way of mounting it that may be more appropriate).
I'm sure that there are other implementations that will suffice, but at
least this one meets with my personal aim of presenting inductors to
the mixer and Tx buffer, and it also allows a diplexer to be
constructed that properly terminates the 2nd mixer.
has used different component
values in his implementation; you can choose. Henning has also used
Wima MKS capacitors in order to avoid the poor behaviour of Hi-K
ceramic capacitors; I have chosen to use 'non-Hi-K' NP0 ceramic
I fitted two 100nF 1206-body 5% NP0 capacitors in parallel, one above
the other, for C651;
these are expensive at 90p each (Farnell 882-0210) but that is not a
major problem in the scheme of things. They have a minimum order
quantity of 10, so fitting two
is also no problem! You could maybe use X7R dielectric at a pinch for
cheapness, but see my
note elsewhere on
Farnell offerings. It is not worth
putting anything worse than NP0 in
any filter or tuned circuit.
The inductor fitted nicely, albeit a bit tricky to position in the
In all of this, I have ignored group delay (as I believe did the
original design). Since we only use a small part of the full bandwith
it is likely but not certain that the effect of group delay will not be
contact me if you would
like to comment on this (or anything else)
This shows the cutoff of the
whole receive path from diplexer input to common-base stage output.
This simulation includes the 'extra' pole at the collector of the
common-base stage (C8 above), which is not really needed.
Below is a magnified view of
the above transmission plot for the complete circuit. The span
15kHz±3kHz has amplitude variation of about 2dB. For SSB we need
3kHz overall bandwidth (±1.5kHz), which is flat to about 1dB.
Well, that filter looks pretty good. Please note that this is with
reduced attenuation in
the PI-pad (120-51-120R). I found that the original 82-100-82R, with
4dB more attenuation, gave adequate signal with the output pot near
minimum, better for achieving low distortion and intermodulation by
running at lower level in the amplifiers.
The extra pole on Receive due to the capacitor across R620 (5k6) is
further 'improving' the slope, but the 'T' LPF is alone is amazingly
The increase to 100nF from 10nF of C638 has maintained a 50R load
except for the onset of the amplitude rolloff, as can be seen by the
slight 'kick' upwards of the input amplitude trace (in green above) at
just over 20kHz. This is difficult to avoid without considerable extra
poles or by changing the original design criteria (possibly relying
entirely on the CODEC filters) and is not a problem.
I think that this is getting close to the best we can achieve without
adding a daughter board. Although it would obviously be inviting to
have a fresh start with the Rx-side 2nd IF, the performance of
this 'low intrusion' diplexer/filter is more than adequate to ensure
correct termination and to provide some alias rejection.
The above schematic represents my modification to the design. I have
been persuaded that it
would be a Good Idea
to increase C638 (C1 on the simulation) from 10nF to 100nF in order to
minimise the 'gap' before the HF arm 'comes in'. This will reduce the
high termination impedance (visible on the simulation plot as an input
amplitude increase) that otherwise occurs between 30kHz and 300kHz. For
of one more change we may as well get it right. This is shown on the
simulation above. It
is best to use another of the NP0 capacitors.
A better input impedance over the full frequency range is obtained with
C638 at 200nF, but because the input becomes shunted by the HF arm at
15kHz, the gain and slope of the transmission characteristic are
adversely affected also. Stick with 100nF!
Coupling Capacitor types
As we all know, Hi-K ceramic capacitors have poor initial tolerance,
temperature coefficient of capacitance, voltage coefficient of
capacitance, and have high losses dependent on frequency. The X7R and
Y5V should not be used in tuned circuits of any sort. Preferred typed
for tuned circuits are NP0 ceramics or, if their self-inductance is
acceptable, plastic capacitors such as polystyrene, polyester or
When used as decoupling capacitors, the characteristics of Hi-K
ceramics are usually tolerable (sometimes paralleled with a smaller
value capacitor better to handle high frequencies), but are they
suitable for coupling signals (d.c. blocking)?
Consider the normal requirement for a coupling capacitor: it must pass
the entire a.c. signal but block the d.c. level. If it actually does
this, then the a.c. signal across it is zero and dielectric type not of
major significance. If it has a significant a.c. signal appearing
across it, then the dielectric characteristics determine whether that
signal will be affected. Perhaps the most worrying of these effects
would be voltage non-linearity of capacitance, where the capacitance
and hence voltage drop would depend on the instantaneous voltage
present across it. This would be manifest as non-linearity of signal
drop with instantaneous amplitude, a source of distortion, which we
clearly do not want!
Since this depends on the voltage across the capacitor, it is clear
that (for a given frequency and circuit impedance) a larger capacitor
value will produce a smaller effect overall than a smaller one, since
it develops less voltage drop and therefore has less overall effect.
If our Hi-K capacitor has a value small enough to allow a significant
a.c. voltage to appear across it, signal distortion is likely. However,
if we use a Hi-K capacitor of sufficient value that any variations of
voltage-drop due to the dielectric are insignificant compared with the
full signal voltage, then not only is the Hi-K effect far less
important, but also as much signal level as possible will be passed to
the subsequent circuit.
The original Picastar uses several 220nF Hi-K coupling capacitors in
nominal 50R paths in the 2nd IF stage, which operates at 15kHz +- 3kHz.
At 15kHz, 220nF has an impedance of 48.23 Ohms. This is the same as the
circuit impedance, so half the signal will be developed across the
capacitor, giving not only signal attenuation but also the likelihood
of distortion due to the dielectric behaviour. This is not good!
I have suggested using 10uF, or at a pinch 2u2. At 15kHz, 10uF has an
impedance of 1.06 Ohms, which is as near zero as we might like. This
means that not only will it hardly attenuate the signal in the 50R
path, but also that any capacitance variation will have only a very
small effect on the signal - it will not give significant distortion.
Clearly, even larger values would be better, but 10uF is a useful value
and is obtainable in small SMT sizes that will fit the pads. Although,
in most cases, the actual terminal voltage difference is small, it is
wise to choose capacitors with 10V rating (this is the supply voltage)
for safety. X7R is the best choice of dielectric, but Y5V might still
not be that bad.
Farnell offers X7R 10uF 16V 1206 10% capacitors (
1463368) for 47p each (singly). The
higher voltage rating of these should provide an excellent margin. A
10V part is also available (1288264) for 19p each, but these are priced
in tens. Either of these would be a good choice.
If it were physically possible, we could use plastic dielectric
capacitors (polystyrene, polyester, polycarbonate) or mica (at a cost).
But values of 10uF are not really practical; we need a large value in
order to avoid the rolloff. Maybe you could consider using these, but
bear in mind that we are trying to improve things, not achieve
perfection. The capacitors mentioned in the previous paragraph will fit
the existing PCB pads and, I believe, are a colossal improvement.
Actual response measurements for the suggested filter have been
performed by others and show good
conformance to the theoretical shape
Henning has built his diplexer/filter with his own chosen values. He
have changed the diplexer after the second mixer to a Tshebychef LP
0.05 dB ripple giving 20 dB return loss in the passband (two times 330
uh, 177 nF (150 nF + 22 nF in parallel; type Wima MKS) to ground in the
These values give about 0.25dB less gain difference across 15kHz +-
1.2kHz, but considerably less attenuation (about 6dB less) at the image
frequency. Because the image and wanted frequencies are only an
octave apart, it is a difficult compromise, but I am happy with the
values I used. Henning has wisely considered my change philosphy and
chosen his own filter preference.
You may prefer to use Henning's values, but remember to use NP0 or MKS
capacitors for stability and low distortion.
On the topic of anti-aliasing, Dave G3SUL
specification requires that
compatible CODECs "have on-chip filtering
incorporated which gives rejection of at least 74 dB at the Rx aliasing
frequencies, and Tx anti-alias rejection of 40 dB for all output noise
from 28.8kHz to 100kHz." Dave suggests that therefore "any additional
anti-aliassing rejection provided by modifications to the 15 kHz LPF
and buffer stage is of no practical consequence to the anti-aliassing
Even though the modifications might not provide any eventual improvement
to the anti-aliasing performance, it is important to bear in mind that
the reasons for changing the filter are:
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- The original Cauer filter topology mis-terminates the 2nd mixer
role of diplexer), so it must be altered to an inductor-input 'filter'
- The second inductor was added principally in order to avoid the
Buffer having a capacitive load, as it originally did - and as a
2-section LC filter still would. This is not quite so
essential after finding defects in this Tx buffer anyway, and providing
a real 50R load for it, but it is generally bad for the stability of a
feedback amplifier to load it capacitively, so the output inductor of
the (now) 'T' filter is still useful. The inductive 'output' arm avoids
the load dropping from 100R to 50R at higher frequencies as capacitor
impedance drops, which is good even for the revised buffer. That
this gives a further pole in
the filter, allowing a better rolloff than a 2-section filter, is a