STAR LO Investigation
In 2015, it has become apparent that constructors can and do end up with too little
or a very distorted LO signal to the Magic Roundabout on 20m only.
This is due to the inclusion in the Matrix of D405, which energises the 2nd (bandpass)
filter on 20m, as well as on higher bands.
This was not the case in Peter's Radcom articles as available on the Group, and wisely so.
Even the nominal filter is on the LF slope when the LO is set to 20m; with worst-case
components it is significantly more so, and if the parts (likely the inductors) are even
lower than expected at 40+MHz, as are many ferrite-cored inductors, then the wanted fundamental can be almost entirely suppressed,
leaving mainly higher harmonics - not at all desirable.
Fortunately it is easily cured by cutting out D405 from the matrix. This will disable
the bandpass filter on 20m and entirely avoid the problem. Because we all use
fast DDS clocks these days together with the AD9951, the residual rubbish is minimised
and is, anyway, likely to be far less trouble than having the filter present!
As always if you have a fully working Star then it would be pointless to
open it up to make the change.
Thanks to Hank, WZ3HMT, for his work on this and for not suffering in silence.
Now, back to the LO stuff!
If you have no time for the background and maths, see
Send comments or criticisms, please!
What constructors have said
(good & bad!)
If you decide to use any of the information here, you will make
your PICASTAR non-standard. Even if you wish to do the changes on a
'new-build' STAR, please build it first using all the standard build
information, with no changes from the published design. Also you need
to calibrate it using all the excellent BASIC programmes, to verify
that it works correctly, before you modify it. This is what I did
If you do not 'RTFM' (read and use all
the PICASTAR documentation in the Yahoo! Groups Pic-a-Project
group before you modify
your build in any way, then no-one will be able to help you to get your
own build working. You must
join this group in order to obtain (and be permitted to use) the design
information that you need. If you then need help (with an un-modified
STAR) you will get it by posting your problem on the picastar-users
If your STAR is modified in any way, then the very helpful people on
the picastar-users group will probably not be able to help you.
Many STAR constructors have suffered from low LO output, which results
the H-Mode mixer drive circuit (squarer) self-oscillating either during
LO signal transitions or, at worst, continuously.
It is generally accepted that a level of +7dBm into the 50R input of
the squarer is needed for satisfactory operation. My LO, built to
the published design on Glenn VK3PE's excellent ComboSTAR PCB, only
produced +3dBm at the high frequency end of range (10m band), so I
looked for solutions to the Yahoo! "homebrew-radios" discussion group.
It soon became apparent that there were a number of fixes in use,
ranging from using a different LO buffer amplifier transistor type, to
replacing the whole amplifier with an AD8008 design! Others had success
On removing an amplifier input-matching resistor and shorting a
decoupled emitter resistor, my STAR LO reached the magic +7dBm at 10m.
On the journey to this I had investigated and analysed the DDS circuit,
filter, amplifier and H-Mode squarer, changing and verifying every
single component, and coming to the conclusion that
nothing in my build was 'wrong'; it just didn't produce enough signal!
I suppose that, having achieved the +7dBm level, I could have accepted
the result and moved on to other areas, but I am a curious person and
felt that there should be a universal solution rather than individual
The LO circuits are shown below as extracts
from the PICASTAR documents Copyright of Peter, G3XJP and drawn by
The official filter and amplifier circuit remains the same as the
original Pic 'N Mix design by Peter G3XJP, but the DDS area has changed
over the years as better ideas and DDS chips arrived.
It may be useful to see the calculations that establish the output
level from the AD9951 DDS circuit. If you are a masochist, look here. If not, just believe that
the transformer primary (6t c.t.) has 1Vpp across the six turns!
Output voltage calculations for the older DDS configurations, using
and AD9851, can be seen here.
I have shown all a.c. measurements as Vpp, since this can be measured
unintrusively with an oscilloscope and is independent of prevailing
impedance. The use of a x10 frequency-compensated probe somewhat limits
the lowest amplitude signal that can be seen on a 'scope, but we are
make big signals, so that limitation only applies when trying to
beyond the filter cutoff.
The current DDS solution for all new constructors uses the Analog
Devices AD9951 device, which has significant improvements over the
previous parts used in STAR, the AD9850 and AD9851. AD9951 uses 1.8V
3.3V internal logic as opposed to the earlier 5V internals, and is much
more easily damaged electrically. The higher permitted input reference
speed of up to 400MHz, together with the 14-bit DAC (10-bits on the
AD9850/51) allow much better
synthesis with reduced spurs.
VK3PE's ComboSTAR PCB has provision for direct assembly of the AD9951
and its peripheral circuit onto the PCB, but it can instead use the
Carrier Board (a plugin sub-module) that is used on the earlier, Pic 'N
Mix based DDS units. In theory this would permit the older, AD9850 or
AD9851-based carriers to be used with ComboSTAR, but it is unlikely to
be a winner due to the lesser performance of these
devices. Most ComboSTAR builders will, like me, build the AD9951
circuit directly onto the main PCB, but some will use an existing
Several people have questioned why the earlier LO had no trouble
providing +7dBm, whilst the current AD9951-based LO struggles. Here is
"Older was Better"
The original 9850 in Pic 'N Mix was used single-ended. It fed a 200R
'source terminating' and d.c. load resistor, to ground because the
current to a ground load rather than sinking current from AVdd
(AD9951). The current waveform of 0 to +10.24mA into the 200R in
parallel with the 200R filter produced 1.048Vpp of signal, with no
transformer losses. Since
the AD9850/51 have +1.5/-0.5V of output compliance, this is not a
This signal feeds the 200R filter, which is practically identical to
that in the AD9850 data-sheet evaluation circuit (the first inductor
value has been increased slightly to 1uH). This gives a reasonable
signal level to the 2N3866 Buffer Amplifier.
The disadvantages of the AD9850
were the limit of 125MHz for the reference clock, making the sin(x)/x
rolloff significant at high output frequencies, and the 10-bit DAC
which limits the output step resolution. The whole 1Vpp output appears
at the filter, with no transformer losses, but sin(x)/x rolloff still
Before long Peter introduced the use of a centre-tapped transformer
rather than directly coupled output. This not only gives higher output
level but also better cancellation of even-harmonic distortion from the
Here things changed. The original single-ended output needed the 200R
load resistor to complete a d.c. path for the DDS output current; the
transformer-coupled version used the transformer primary to provide the
d.c. path. As with the Analog Devices example circuit, the DDS output
now uses no resistors, the a.c. load being the filter and subsequent
termination, and the d.c. path being the transformer winding. No source
termination means higher output signal, since no
power is lost in source terminating resistors.
In fact, the output from this version is nominally 1.5Vpp into the
Newer is better!
Then the super AD9951 arrived on the scene. The output circuit was
provided by the Analog Devices Applications team after the STAR team
had had little success at adapting the previous designs. Being a low
voltage, small geometry device, the AD9951 cannot withstand any abuse
such as overvolting the outputs. The previous transformer-only coupling
could easily damage the device; with no load or with the increases in
load impedance above 200R caused at higher frequencies by the filter
characteristics, the device would exceed the +-0.5V output compliance
limit and not like it.
The AD recommended 9951 circuit puts 100R directly across each
half-primary of the output transformer. This limits the DDS output
excursion to +-0.5V even with no filter connected (barring commutation
spikes due to transformer leakage inductance) and produces 1Vpp across
the full primary when loaded with 200R by the filter and amplifier.
This DDS naturally became the part of choice for
builds; it has far superior performance, and a replacement DDS carrier
plugin was designed to give easy upgrade on existing Pic 'N Mix based
STARs. With a 3+3t:6t transformer, the output is nominally the same
1Vpp that appears across the primary. This feeds into the DDS
reconstruction filter as before.
Because of the re-introduction of source termination at the DDS with
design, the output to the filter reduced from 1.5Vpp using AD9851
to 1Vpp, both nominal and subject to sin(x)/x reduction of amplitude at
high frequencies. The net result of this is that the input to the
amplifier is considerably lower than with
the earlier transformer-coupled designs.
The single transistor
amplifier circuit is shown below using the component references from
The Amplifier is a conventional single-transistor stage using a 2N3866.
It is a clone of the design which originally appeared in QST of
1981, Hayward & Lawson's "Progressive Communications Receiver",
with identical topology and component values.
It has a low input impedance at the base, and is transformer fed at the
the 56R load on the MR.
The 2N3866 has a current-gain bandwidth product fT
of 800MHz and minimum hFE of 25; at our
top LO frequency of 40.5MHz the reduction of gain due to fT
is dominant (at 40MHz the current gain becomes about 20 regardless of
current gain). The
input amplitude has already reduced from the 1Vpp at the DDS outputs by
sin(x)/x rolloff, now we also have reducing current gain due to the
800MHz fT of the 2N3866. Oddly, this was
somewhat in the AD9850/9851 transformer-fed implementation because of
an imperfection in the filter - the input
rises well above
200R between 26MHz and 50MHz, peaking at about 330R at 31MHz and then
reducing somewhat again. This removes some load from the DDS, whose
correspondingly increases and counteracts the reductions in amplitude.
This is less pronounced with AD9951 because of the source termination,
without which the DDS chip might be (and may have been) destroyed as
The Amplifier is fed from the filter via a series 180R resistor, R3311.
This resistor pads the low input impedance of the amplifier input
and acts as a good terminator for the Filter. Unfortunately, most of
the signal is lost in this resistor. Taking the amplifier input Z as
approximately 20R, shown by measurement and simulation, the potential
division is large. If all of a 1Vpp DDS output reaches the output of
the filter, the termination reduces it to only 0.1Vpp at the amplifier
input. The amplifier has to work hard to achieve the required output of
+7dBm into 50R, which is 1.42Vpp. Bearing in mind that we started off
with 1Vpp (for AD9951), it seems odd now
to amplify back up to pretty well the same voltage to overcome the loss
in a matching terminator! Of course, we are also changing the impedance
to 50R for the 'Magic Roundabout' (MR) squarer.
When using the AD9850/9851 DDS with transformer output, there was a
amount of drive available (around 1.5Vpp), so the Amplifier could
relatively comfortably achieve over +7dBm output. With the AD9951 DDS,
the lower level of 1Vpp, coupled with the small increase from filter
input impedance rise and the loss from sin(x)/x loss, has left many
constructors short of output from the amplifier. What can we do about
Conventionally the first recourse is to remove the 56R (R312 on
which is shunting some base drive from the amplifier for no good
reason. The amplifier input impedance is only 20-30R itself, not high
impedance, so the resistor is nugatory and serves only to reduce the
gain. Some people, me included, still did not achieve
+7dBm. A cure for this has been proposed: to use a 2N5109 in place of
the 2N3866. 2N5109 has an fT of 1.2GHz,
higher than 2N3866, which together with a higher minimum gain makes it
able to provide more amplification. This said, I know of one case
where 2N3866 gave significantly better output than 2N5109!
But this is only tweaking things. It is quite disconcerting for new
builders, just STARting out, to find that they need to adjust this,
change that, and so on, in order to achieve the performance that seemed
so easy for others. Maybe foolishly, I took up the challenge to find a
There may be other solutions; Robert G3WKU has suggested increasing the
Amplifier input impedance so that less signal need be lost in the
filter termination, and others have simply replaced the
Amplifier with an IC version using AD8008. G3WKU currently uses a
2:1 turns ratio transformer to match better the 200R filter output to
the low input impedance of the amplifier and avoid the resistive loss
in the present scheme.
I have unilaterally decided that the optimal way for most STAR builders
is one which does not
require any extra components on the existing physical layout. Any
use straightforward components similar to the existing design and need
be compatible with use of the AD9951 DDS device.
Although this cannot yield results as good as a complete redesign of
the LO (both electrically and physically), for most of us the need is
simply to obtain good results using the existing PCB layout and with
minimim disruption. For STAR builders who use the original, modular
style of construction, it might no be so difficult to replace the whole
DDS/Filter/Amp arrangement with something else, but ComboSTAR builders
will probably prefer a less intrusive solution even if it does not
Best Possible Performance. Any practical design is a series of
compromises; in this case the re-use of an existing physical
layout. As Voltaire once said, "The best is the enemy
of the good". He probably wasn't thinking specifically about STAR, but
you never know - he might have been.
The answer for us is to reduce the filter impedance from 200R.
This is almost trivial at the DDS end; it is only necessary to wind
fewer secondary turns on the transformer (or unwind a few if an
existing transformer has been constructed as suggested, with the
secondary on top). At the Amplifier end the only changes, with luck,
will be component values to reduce
As a later comment, I have
become painfully aware of the poor performance at even medium
frequencies of the type 43 ferrite material (see my comments relating
to the bidirectional amplifier).
This material is very lossy at 10MHz, and the Al reduces between 1MHz
and 10MHz until it is little different to the far less lossy type 61
ferrite. Since here the LO uses high-side injection, the LO frequency
is always above the IF (10.7MHz normally) so there is great benefit in
using a BN61-2402 core for the DDS transformer. Also, multi-filar
winding causes a transformer to behave as a transmission-line rather than flux-coupled transformer at higher
frequencies. This causes far less flux to be coupled by the ferrite,
further reducing the effect of core losses. Anyone winding transformers
for the LO would do well to use a BN61-2402 core with trifilar winding
for the differential to single-ended trasnformation at the AD9951
output, and to ensure that the DDS Buffer amp output transformer is
also wound bifilarly and preferably on a material 61 ferrite core. This
will maintain good coupling and reduce losses, especially at high
frequencies where it is so necessary.
Unfortunately low impedance filters have worse characteristics in
general than high impedance filters, so obtaining the relatively sharp
cutoff of the original filter is impossible if we are to have
reasonable in-band characteristics (which we need).
Fortunately, we have opportunities now that were not possible with the
original design; to stick with a problem that we can avoid just because
of history is not a good idea. The AD9951 got us into this position; it
will also allow a remedy.
Remember that the original AD9850 had a maximum clock input of 125MHz,
putting the first image frequency fairly close to the wanted frequency
at the higher frequencies and necessitating a sharp filter cutoff. The
AD9951 will accept up to a 400MHz clock (sample rate) and the Butler
oscillator will easily allow use of a crystal capable of providing in
excess of 150MHz - the crystals are readily available and there is
absolutely no reason not to use 150MHz or so, or even higher,
especially on a new build.
Because of this, the first image frequency is much higher, leaving a
larger frequency 'gap' into which we can fit the filter cut-off. At
150MHz clock the first image is at 150-41MHz, 109MHz, a 'gap' of 60MHz
in which to roll-off the filter! Using even the highest permitted clock
on the earlier AD9850 device (125MHz), the image is at 84MHz, leaving
far less margin for tolerances and filter slope.
To make best use of this opportunity, we have to choose 'the right'
filter impedance. There are a few things to consider, apart from
whether we can actually implement a filter!
- Neither the filter passband 'ripple'
in amplitude, group delay and phase, nor the out-of-band
characteristics, is an important consideration as they would be if the
filter were reconstructing a complex signal; we have only static
frequencies to consider. The Cauer filter as originally used is well
suited since it rapidly transitions between passband and stopband.
- The filter impedance needs to be low enough that any
at the Amplifier does not throw away too much signal. We already saw
that the 200R circuit loses 90% of the amplitude in the translation to
the amplifier input impedance, so we can accept that Lower is Better.
The chosen impedance should be available by reducing the secondary
turns of the DDS output balun, TT1. This is straightforward and avoids
major change to the transformer design, which is known to work. This
gives clearly defined step values.
We must be able to implement a reasonable filter, with lowish in-band
amplitude and impedance ripple and good out-of-band attenuation. It
would also be nice to obtain re-entrant out-of-band attenuation of
<50dB for images, if we can, to improve on the original filter!
impedance variation with frequency can reduce output significantly if
it drops; the original rising impedance approaching cutoff may account
for early destruction of AD9951 devices used without the present
source-terminating resistors, but is less troublesome with the present
It is fully reasonable to demand a DDS sample rate (Butler clock
output) of > 150MHz with present technology. Almost everyone is now
using clocks of this specification; there is every reason to do so and
no good reason not to. The sin(x)/x high frequency loss of signal
reduces as well. It's all win, no lose!
Initially I considered a filter of 139R. This is what you get with a
3+3:5t balun istead of 3+3:6t, one less secondary turn.
The filter designs quite easily with this impedance. Unfortunately the
benefit from the reduced terminating resistor is rather small, also
considering that the signal amplitude from the DDS has reduced. Instead
of losing 90% of the signal voltage in the terminator; ending up with
0.1 x the filter output voltage at the Amplifier, we get 0.143 of the
filter output voltage. But the DDS output voltage, and hence the filter
output, have reduced because of the 5t rather than 6t secondary; from
1Vpp to 0.833Vpp. The net result is that the voltage presented to
the amplifier input changes from 0.1Vpp to 0.119Vpp, not an
earth-shattering increase and not quite the answer.
I skipped the next step of removing 2 secondary turns, and decided to
see whether it was possible to design a reasonable filter to suit the
removal of 3 turns, making the transformer 3+3:3t. This would also be
ideal for trifilar winding to increase coupling (reducing leakage
inductance netween prinary & secondary), which should be of
benefit, and results in the need for a filter of 50R impedance.
Having learned my lesson from the first attempt, I resolved to
determine the benefit for amplifier input level before trying to design and
construct a filter. The DDS output voltage is now halved, 0.5Vpp
nominal. The padding resistor at the amplifier input needs to be around
30R to give a reasonable termination of 50R with the amplifier input
impedance of approximately 20R. The signal arriving at the amplifier
input is therefore 0.5Vpp x 20/50, which is 0.2Vpp, double what we had
with 200R and hopefully more than enough to avoid low output problems!
The filter design requirement was for an initial response that is level
in amplitude up to 42MHz then falling off to -60dB somewhere in the gap
before the first image frequency at 84MHz and hopefully remaining near
or below -60dB thereafter, to achieve an improvement over the original
filter for high images (the original filter rises to worse than -50dB
on these images).
I used the excellent Elsie
student edition analysis and design programme from James L. Tonne,
without which the design and characterisation of the filter would be
very tedious! I have taken the liberty of including some of Elsie's
This is the new, 50R filter design for simulation and with normalised
The result of simulating the above circuit shows no enormous
transmission disturbances within or outside the passband. The input
impedance fluctuates within the passband, but so did the original design.
For comparison, I have included the simulation of the original STAR
200R filter. This is the circuit
The simulation of the original STAR
filter shows impedance ripple in
the passband together with disappointing rejection of the higher
images. The filter following the Buffer Amplifier further removes the
Original STAR Filter, 200R
Having obtained a reasonable filter in spite of the low impedance, I
altered the components to preferred values in the normal way, and
checked that the inductors were reasonable to wind. The circuit and
simulation above use these values, so they are representative.
An advantage of the final filter is that the capacitors are all of
larger values than their counterparts in the 200R filter, so stray
capacitance is less significant. The inductors are not the horrifyingly
low value that I expected, so can easily be wound, and track lengths
will not upset them. I then modified my STAR to use the new balun and
filter components and checked the results.
With the original circuit, in order to achieve +7dBm, I had removed
the 56R base shunt, R312 (ComboBoard reference) and completely shunted
the emitter 5R6 (R316) in order to scrape +7dBm output at 40MHz in
'sig-gen' mode. I didn't change the 2N3866 to 2N5109; I didn't have a
2N5109 and this seemed quite uninviting to try, although it may have
After changing the balun and filter, without restoring the original
amplifier circuit, the amplifier happily delivered over 2.25Vpp, in
excess of +11dBm right up to the filter cutoff frequency of 45MHz, with
+13.5dBm (3Vpp) at lower frequencies. The 3Vpp output is actually too
high (!); it corresponds to a 6Vpp signal on the collector, but the
static transistor c-e voltage is only about 7V so the transistor is in
danger of clipping peaks if there is a slight amplitude increase for
With gain to spare, the 'problem' remaining is to reduce it in order to
prevent the Amplifier overloading at any frequency! It's rather easier
to lose gain than to -um- gain gain. Firstly I re-instated the emitter
degeneration resistor that I had shorted before to increase the gain.
The value actually 'jumped' to 10R, partly because of contemplating
binary search for a good value, partly because my original build used
two piggy-backed 10R in lieu of 5R6, afterwards further piggy-backed
with a 0R to
increase the gain - when I unsoldered the 0R, one of the 10Rs came with
This degeneration gives local -ve feedback in the circuit, in addition
to the overall -ve feedback through the 1k resistor from output to
base. The benefit is that variations in the circuit (the transistor
itself, mainly) are levelled a lot, making the gain less dependent on
the individual transistor. It also raises the input impedance, but this
is mainly affected by the 1k overall feedback resistor.
and maybe desirable is to adopt W4ZCB's suggestion of reducing the
capacitor on the emitter (10nF) thereby increasing the emitter-current
feedback at low frequencies. This is worth bearing in mind if the
output is Too High at lower frequencies (too high! That would be a
treat!). It may also be useful to reinstate the input shunt
resistor R312 (was 56R) in order to pad down the input impedance of the
amplifier if the emitter resistor is less decoupled; it will reduce
the overall gain but make the amplifier input impedance and filter
termination impedance less dependent on individual transistor
characteristics. If you find these things necessary I would be pleased to hear from you.
The response with the emitter degeneration back in place, but with the
resistor doubled to 10R is very promising. I measured the actual levels
on my unit from 20MHz to above the high cut-off; because the second
filter (following the Buffer Amp) stays active below its cutoff
frequency in sig-gen mode (TrxAVR
does this, probably Pic 'N Mix is the same), the results at and below
20MHz were not completely useful, but this is below the 20m band (14MHz
+ 10.7MHz IF is 24.7MHz). On lower bands the 2nd filter is switched out
in normal use, so this is not a concern.
All signal levels are Vpp, measured on a
100MHz oscilloscope with a
properly compensated 100MHz 'x10' probe.
R312 is omitted, see note in parts list.
||DDS-OUT of balun
|Filter out (50R)
| Buffer Amp I/P (base)
|| Buffer Amp O/P (50R)
|Squarer I/P (50R)
||Output Level (dBm)
1) In 'signal generator' mode, the secondary band-pass filter
the Amplifier is activated when normally it would not be at this LO
frequency. The signal is below the filter pass-band, so the Buffer Amp
is not properly terminated and no ouput reaches the Squarer Input.
2) At 25MHz, where the 2nd filter is beginning to cut response, the
squarer input is 1.5Vpp, +7.5dBm, but the amplifier output is higher at
2Vpp, +10dBm. The 2nd filter may be presenting a load greater than 50R,
which would cause this.
3) At 30MHz up to 40MHz, corresponding roughly to 20MHz to 30MHz at RF,
the 2nd filter is in its normal passband and the output to the squarer
is from 1.8Vpp to 2.4Vpp, corresponding to +10 to +11.6dBm at 50R.
4) At 40MHz the amplifier output is 2.0V and the 2nd filter 1.8V,
suggesting that in fact the 2nd filter is beginning to bite.
5) At 45MHz the DDS filter output is just holding up, but the 2nd
filter is cutting output very noticeably, probably mis-terminating the
amplifier which now produces 1.7Vpp output, still +8.6dBm!
6) From 50MHz upwards the DDS LPF rapidy reduces amplitude at its
output (and of couse the Amplifier input). The input and output of the
amplifier have become negligible at 65MHz (15mVpp from the amp output
is -33dBm) and zero by 70MHz, nothing visible at all on the 'scope.
This corresponds well with the design characteristic of the filter! and
The considerably improved output at the high frequency begs the
question of whether the circuit would stand use at 6m, with an LO
frequency of (52 + 10.7) = 63MHz. Although this was not within my
original design aim, I believe that it would. I'm tempted to try it;
the DDS output holds up well up to at least 70MHz and the amplifier has
'gain to spare'. Anyone wishing to try this would be advised to use a
higher reference oscillator frequency than 150MHz, to push the first
image frequency higher and make the filter less critical. The Butler is
quite capable of 200MHz with a 'hairpin' inductor in place
of the coil, but my own crystal was initially unwilling to work
reliably at higher than its
stated frequency (152 and a bit MHz). Robert, G3WKU, reports success
with a 40MHz fundamental-mode crystal oscillating at 5th harmonic using
a hair-pin 'coil'. M0RJD has now achieved 213MHz by selecting the 7th
overtone of the 5th overtone crystal, by changing the damping resistor
R304 to 1k. A separate oscillator could be
used, as some are
already doing. The AD9951 has 400MHz as the highest specified
If limited to a 150MHz reference, it is probably not worth trying to
achieve 6m. The LO output needs to be 63MHz for 'high LO', so a filter
corner frequency of
68MHz would be reasonable. The first image, which we must reject
deeply, is at (150 - 63)MHz = 87MHz, so there is not much margin (we
would aim for -60dB at say 80MHz or less, which is a bit tight and may
not be fully effective when considering tolerances in the
filter!). The required slope of the filter is also steep and may result
in a poor response for in-band signals. The input impedance of the
filter seems most badly affected with this filter architecture when
trying to achieve a high slope, especially at 50R; don't neglect
it! A better solution for 6m is to use a considerably higher
reference frequency. 175MHz should be achievable easily by the
Butler circuit, if a suitable crystal can be found; it pushes the first
up to 112 MHz from 87MHz and gives much more filter design leeway, as
well as reducing the sin(x)/x rolloff inherent in the DDS digital
sampling. For higher Butler Oscillator frequencies, the resonant
frequency of L301 in the tank circuit must be increased, so less turns
(or even a 'hairpin') will be needed.
To put this into perspective, all these things are equally important
regardless of how 6m is achieved. In fact, everyone so far that I know
of who has extended STAR to 6m has also replaced the Amplifier with
another design based on AD8008 in order to get adequate output - this
may no longer be necessary, especially if the 2N3866 is replaced
with a 2N5109 with higher fT.
I have no intention of stretching my STAR to 6m in the immediate
future, so my filter design is only good for the standard
STAR bands, but if anyone does this using a 50R filter I would love to
hear from them.
My recommendation to anyone newly building STAR is to use the method I
have described in order to greatly improve the LO output level. Others
may not wish to rebuild the DDS filter on a working STAR; it doesn't
make sense to do so (if it ain't broke don't fix it). The changes
affect only component values; no extra parts are needed and every
component already has a place on the existing layout. The increase in
LO output easily exceeds the target +7dBm, even having takes steps to
reduce it (a thing that not a lot of recent STAR builders need to
do!). I have shown why the earlier AD9850/9851 designs gave more
LO output due to the DDS output architecture having no source matching,
and explained why this architecture is not safe with the AD9951. The
changes to the filter easily improve the LO output and are in no way a
disadvantage to anyone using the AD9951. The new filter is neither
needed nor recommended for use with the earlier DDS (AD9850) which has
a maximum clock rate of 125MHz. AD9951 users could increase the clock
to as much as 180MHz (5V supply), which would allow the new filter to
be used, but it already provides higher output and
the change is not required. New builders will automatically use AD9951,
if only because of its better DAC (14-bit rather than 10-bit) and
should seriously consider using the 50R filter in order to completely
avoid LO amplitude problems.
Of course, if you are unlucky and do not get any benefit from this
filter change, then although I will listen to what you say I can't do
any more than that. Your mileage may vary. But do let me know what results you get!
I am grateful to Glenn VK3PE, Duncan G4ELJ and many others who
have spurred me towards this solution, and especially Robert G3WKU who
has checked the calculations and generally joined in with the design
and analysis process. Also profound thanks to Peter G3XJP, the 'father'
of PIC-A-STAR. Without his generosity we would not have the STAR in the
Please remember that the above is Not
an Official Modification, but do it just the same!
Now some of you will want to know the filter details.
The filter has a sharp cut-off starting at about 47MHz and reaching
-60dB at 75MHz, which easily chops off the first image frequency of the
DDS output. Above 75MHz, the attenuation worsens to 58dB at some
For comparison, the original 200R filter cutoff begins at 43MHz and
first reaches -60dB at 65MHz, worsening thereafter to as little
as -45dB at higher image frequencies.
The slower cutoff of the new 50R filter is necessary in order to reduce
in-band effects, most notably a severe drop in input impedance. Since
the first image with 150MHz clock is at 109.5MHz or above, this is very
comfortable. Even the AD9950 with 125MHz clock has a lowest first image
of 84.5MHz, still 10MHz in hand.
The greater stop-band attenuation of the new filter is another bonus,
since higher images will be attenuated by at least an extra 10dB.
These are the normalised, preferred values for capacitors and realistic
targets for winding the inductors.
When winding inductors, I
found that measurement showed a higher
inductance than the calculated number of turns suggested. The strong
recommendation is to
measure your inductors and adjust the turns for the specified values;
there are not many turns and it is easy to start with the calculated
number and remove turns one at a time if necessary until close to the
design values. I have given the turns I used on T-25-10 cores, but
T-25-2 cores are probably
good if the turns are adjusted appropriately. I wound my
inductors with 26awg Tefzel insulated wire-wrapping wire, which is
tough, easy to use, and fits, but almost anything that will fit will
Large diameter wire will ensure a high Q.
I have used the component references from Glenn's excellent ComboStar
PCB design, since this is what I have built, and included the
used in the original 200R filter for cross-reference:
Parts List for 50R Filter
|160pF if available
|(9t 26awg wire-wrap wire on
T25-10), (8t on T25-2 by SP5TAA)
|(8t 26awg wire-wrap wire on
T25-10), (7t on T25-2 by SP5TAA)
|I omitted this but it will aid
correct filter termination & give some attenuation.
|BN-43-2402, bifilar primary,
This is an edited version of Glenn's ComboStar circuit, to show the new
Reminder for new
When winding toroids, remember the rule that 'each time the wire passes through the
centre it is one turn'.
If the toroid is wound with the wire entering from one side of the
centre hole and exiting from the other side, to make an inductor or
transformer that stands vertically, this is very important and not
intuitive; there is probably one more turn than it seems.
At first glance, the toroids shown in
the photo appear to be wrongly labelled, but the one on the left has
two passes through the centre and that on the right has only one.
In the photo of my board above, you
will have noticed that the Amp output transformer toroid L3303 has more
turns than specified. This was as a result of early attempts at
increasing output level, trying less and more turns. It made no
difference, so I left it at 'more' turns so as not to remove &
replace it excessively. In both photos, the flux residue from a million
component changes can be seen, as yet uncleaned! Heatsinks will be
fitted. I have also fitted a 68R for R312 in order to pad the Amp
input impedance. It reduces output amplitude from 2.0Vpp to 1.7Vpp at
40MHz, still an output of +8.6dBm. The
Since the measurements, my Butler Reference Oscillator
is now running satisfactorily at the 7th overtone of the crystal rather
than the 5th. This now provides approximately a 213MHz reference clock
to the DDS, which runs slightly warm to the touch at 40MHz output. The
inductor is a hairpin of wire-wrap wire in the photo, but is now a
similar piece of heavier-gauge wire for mechanical stability. The
damping resistor R304 is 1k. The trimmer is set to
a very slightly lower frequency than the crystal would suggest, in
order that it starts up reliably. This increased reference
frequency would easily
permit LO use at 6m, with a re-redesigned filter...
The 152.291667MHz Crystal is from 'anonalouise' on UK eBay
and cost £5.00 with £2.00 delivery. It has a rather poor
temperature drift, but once the oven has stabilised it is OK. The
crystal has 3 leads. The centre lead, which is welded
to the can, I bent up along the can and it is not connected to ground.
The can is in contact with the tab of the heater transistor TR304 which
is decoupled to ground; it seems not to be troublesome. In the picture,
the tab of TR304 is not cropped; this is because my original Xtal was
in a larger can.
oscillator running at 213MHz 7th overtone
The blue wires are the yet-uncropped leads of the thermistor.
The orange wire is the tank 'coil'.
Send comments or criticisms
Is anyone actually
I will summarise the information that constructors provide, together
with relevant comments. I have the authors' permissions to publish
No permission, not published.
Robert G3WKU helped me through
some calculations and kept my enthusiasm going! He has not used the 50R
filter because his STAR is fully operational - but this is with his own
modifications to the DDS Filter/buffer.
Robert says "I think it fair to say that you have well and truly
'nailed down' this problem once and for all."
Thank you for your considerable and welcomed help, Robert.
SP5TAA has modified his filter as described, but using a
trifilar-wound DDS output transformer and T25-2
cores (7t & 8t) with excellent results. He gets +9.2dBm at 29.7MHz
with 10.695MHz IF. It would be easy to increase this level if
The cutoff is between 42MHz and 45MHz, with -34.5dBm
Witek says "I think that your modifications are what all STARters
should use in their builds."
Steve G7WAS is now using the
filter with trifilar DDS balun in order to overcome persistently low
output and is now
considering how to reduce
output in order to avoid overloading the
Buffer Amp! One idea is to use a splitter to provide some signal to a
fron-panel connector to enable the DDS to be used as a signal
generator. Steve is also investigating redesigning the filter for
75MHz. His PCB has suffered damage from a succession of 'amplitude
improvement' mods. Previously, Steve found that changing from 2N3866 to
actually reduced the output. He says "When I’ve had problems, with my
Star build in the past, I found that it’s very easy to think it’s just
me; everybody’s build works as designed. It becomes a bit
disheartening, I find myself picking up little clues from various
sources, that all’s not what it seems; others are having the same
problem and have come up with all sorts of workarounds."
Dudley NO5L has measured
the response of his build of the 50R filter using his N2PK VNA and says
"It is flat with a 0.5db valley at about 25Mhz up to 42Mhz. Then
it falls to about -60db at 55Mhz and then rises up again towards 60Mhz;
I can't check it above that. But I'm sure it is as good as the
prev. iteration of filtering, and I don't think I need to fiddle with
things since the pass band is excellent below 42Mhz."